Phase modulation receiving system



Dec. 25, 1945. B. TREvoR PHSEMODULATION RECEIVING SYSTEM Filed June 24,1943 2 Sheets-Sheet 1 imma/95g INVEN-roR 5f@ M24/w 7km/0f?.

BY ATTORNEY mPEwQ Dec. 25, 1945. B. TRE-:VOR

PHASE MODULATION RECEIVING SYSTEM 2 Sheets-Sheet 2 Filed June 24, 194:5

H.. 0 V Y wm i M E MT an N WM .n H 5m m 5 w c o m m 2 0 n .f 0 W APatented Dec. 25, l45

2,391,807` PHASE MODULATION RECEIVING SYSTEM Bertramy Trevor, Riverhead,N. Y., assignor toA Radio Corporation of America, afcorporati'on of"Delaware.

Application June 24, 1943,v SerialiNo. 492,037

Claims.

My present invention relates to radio receiving systems adapted forreceiving phase modulated carrier wave energy, and more particularly toa frequency modulation receiving system particularly designed forreceiving phase modulated carrier waves.

As is well known, phase modulated carrier wavesare generated in a mannersimilar to the generation of frequency modulated carrier wave energy inthat in both phase and frequency modulation, a predetermined carrier.frequency is deviated in response to variations in amplitude of themodulation signals, andthe rate of frequency deviation is a function ofthe modulation frequencies. In the case of phase modulation the highermodulation frequencies are emphasized more than in frequency modulation.Assuming that theY modulation signals are of audio frequency, in phasemodulation there is a greater frequency deviation of thefcarrier for thehigher audio frequencies than for the lower frequencies, and thisfrequency deviation is proportional to the modulating frequency.

At the receiving system, therefore, the audio amplifier frequencyresponse isr deliberately de'- emphasized for the upper audiofrequencies where phase modulated carrier wave energy is received withla frequency modulation receiver. Indeed, in past methods of receivingphase modulated carrier waves it has been customary' to utilizede-emphasis proportional to the audio frequency in the audio network ofthe receiving system. The overall resultr of such de-emphasis at thereceiver is to provide for the combined transmitter and receiver systemsa substantially flat audio response down to the lowest frequency coveredby the de-emphasis circuit. Due to the nature of the cle-emphasiscircuit, however, a loss of gain of about 5:1 is obtained even at thelowest audio frequencies employed.

It may, therefore, :be stated that it is one' of the main objects of mypresent invention to prof vide a frequency modulation receiver capableof receiving phase modulated carrier wave energy, the modulation signalutilization network being constructed and arranged so as to provide anoutput which is inversely proportional to the modulating frequency, butwherein thereis substantially no loss ofy gain for the lowest usefulmodulation frequencies.

Another important object of this invention is to provide a receiver ofangle modulated carrier wave energy whose modulation signal utilizationnetwork is provided with at least two cascaded" resistance-capacitancecompensation networks;

one of the. compensation networks .bei-ng adapted to reduce the highermodulation frequency re.V sponse, while the other compensation networkfunctions to emphasize the lower modulation frequency responseY which isnot aiected .by the first-named compensation network.

Still another object of my invention is to provide an audio frequencyamplifier network which has a first resistor-capacitor network`functioning to cause the high frequency response to be inA verselyproportional to frequency, anda second resistor-capacitor network actingto render the low frequency response inversely proportionalv tofrequency down to the lowest modulation fre` e'ciency ofr frequencymodulated receivers whenused' to receive phase modulated signals, andmore especially to provide compensated audio networks for such receiversin an economical and reliable manner.

The novel features which I believe to tbe characteristic of my inventionare set forthwith particularity in .the appended claims; the inventionitself, however, as to both its organization and method of operationwill 'best be understood by reference to the following description,taken inconnection with the drawings, in which I have indicateddiagrammatically several circuit organizations whereby my inventionmaybe carried into eect.

In the drawings-r Fig. 1 is a circuit diagram of one embodiment n of theinvention,

generic term angle modulated carrier wave is to be understood as broadlyincluding frequency modulated, or phase modulated, carrier wave energy.Those skilled in the art are fully aware of the manner of constructing afrequency modulation (FM for brevity hereinafter) receiver. Therefore,it is not believed necessary to disclose the various transmissionnetworks which precede the FM detector I. Reference is made to U. S.Patent No. 2,121,103 granted June 21, 1938 to S. W. Seeley, disclosingand claiming a. wellknown form of FM detector circuit that may beutilized to derive the modulation signals from FM wave energy.

It is known that the FM detector i may be immediately preceded by anamplitude modulation limiter stage. Furthermore, let it be assumed thatthe various selector networksY of the receiver are designed to pass FMwaves having an overall frequency deviation of at least 150 kilocycles(kc). Of course, the invention is not limited to this specic deviationrange, since it is equally applicable to various frequency deviationranges. Those skilled in the art are fully aware of the fact thatcertain systems of transmitting FM wave energy deliberatelypre-emphasize the higher modulation frequencies by causing the frequencydeviation of the carrier to be greater at the higher modulationfrequencies. In phase modulated (PM for brevity hereinafter) systems thepre-emphasis of the higher modulation frequencies inherently takesplace.

Accordingly, in the prior art it has been the usual practice to providea series resistor-condenser network across the output load of the FMdetector circuit when phase modulated signals are to be received with afrequency modulation receiver. This resistor-condenser networkfunctioned to compensate for the pre-emphasis at the transmitter. Thecompensation network actually discriminated against the highermodulationfrequencies, and was for this reason referred to asde-emphasis. Reference to Fig. 2 shows the pertinent idealizedcharacteristics of prior FM receivers, and it will be observed that theeffect of the pre-emphasis and de-emphasis characteristics was toprovide a compensated' output. In order for thel de-emphasis circuit togive proper compensation, the series resistance should be at least livetimes the reactance of the condenser even at the lowest useable audiofrequency. This gave a 5:1 loss of gain at this lowest frequency.

According to my invention the single de-emphasis network is replaced bya pair of cascaded compensation networks. One of these networks providesthe characteristic shown by the solid line Rz-Cz in Fig. 3. The othercompensation network provides the overlapping characteristic representedby the dotted line R1-C1 in Fig. 3. (The dash symbol between the R and Cletters employed above and hereinafter is not a minus sign, but merelysignifies the physical combination of the respective resistance andcapacity.) The overall effect of these two compensation networks is toeliminate the loss of gain at the lower audio frequencies, and stillhave the audio output inversely proportional to modulating frequency. Inother'words, one of the compensation networks reduces the highmodulation -frequency response in the usual manner, while the othercompensation network emphasizes the lower modulation frequency responsein such a manner as to give a net response inversely proportional tofrequency.

lIIS

Considering the specific circuit elements for accomplishing theaforesaid functions, reference is made to resistor R1 which is arrangedin series with condenser C1 across the detector output terminals. Theconstants of these series resistor and capacitor elements are so chosenthat there results a marked decrease of higher modulation frequencyresponse. This is indicated by the dotted line which is referred to asRi--Ci in Fig. 3. The control grid 2 of the following audio amplifiertube 2 is connected to the junction of resistor R1 and condenser C1. Theaudio amplifier 2 is of the usual construction, and specifically employsa tube of the pentode type. The screen grid of the tube is connected tothe positive terminal of the direct current source (not shown) througha, resistor 3. The anode, or plate, of tube 2 is connected to thepositive terminal +B of a suitable direct current source through a pairof resistors R2 and Ra arranged in series. The lower end of resistor R2is connected to ground through condenser C2. The grid 2 requires adirectI current return resistor to ground. rIhis would normally beincluded in the FM detector load circuit. If this is not done, a gridleak resistor from grid 2 to ground is required. In that case the leakresistor value would have to be at least five times the reactance ofcondenser C1 at the lowest compensation frequency of the network.

The series resistor-condenser network Re-Cz acts as the secondcompensation network. This compensation network has the responsecharacteristic indicated by the solid line referred to as Rz-Cz in Fig.3. The network R1-C1 causes the higher modulation frequency response tobe inversely proportional to frequency. The compensation network Rz-Cz,however, causes the lower modulation frequency response to be inverselyproportional to frequency down to the frequency where the ohmicimpedance, i. e., resistance, of resistor R3 remains greater than aboutfive times the reactive impedance of condenser Cz, It is assumed thatthe screen and cathode bypass capacitors of tube 2' are of suitable sizefor the audio frequency band used.

Generally speaking, the product of R1 and C1 should be equal to theproduct of R2 and C2 on the assumption that R3 is much larger than thereactance of condenser C2, and the impedance of the compensation networkRz-Cz is much less than the plate resistance of tube 2. In Fig. 3 thereis, also, shown the overall response or Corrected audio output curve,and by way of contrast the Uncorrected audio output curve. In the priorart using the compensation network R1-C1 it is required that theresistance Ri be at least ve times greater than the reactive irnpedanceof condenser C1 at the lowest useful audio frequency. This requirementmust be fulfilled in order that the compensation be correct for allaudio frequencies above the lowest used.

This arrangement thereby inherently gives a five-to-one loss in gain atthis lowest frequency. In the present invention, referring to Fig. 3,the compensation circuit R1-C1 meets this 5:1 impedance ratio at afrequency of approximately 1,000 cycles, which is considerably above thelowest useful frequency of 400 cycles. Hence, the loss in gain of 5:1occurs at about 1,000 cycles, and is much less than this at the lowerlimit of 400 cycles. In other words, this circuit does not properlycompensate the range below 1,000 cycles. In order to effect compensationin this range, the second compensation circuit aaah-cov in the plate oftube 2 is supplied. This network operating in the plate circuit of tube.2' serves to raise the gainV of the audio frequencies Vbetween 1,000and 400 cycles in such a manner as .to effect proper compensation. Inother words, R1--C1 de-emphasizes the higher audio frequencies, whereasRz--Cz emphasizes the lower audio frequencies in `such a manner as .togive the desired compensation.

In theV curve sheet shown in Fig. 3, the net compensation response curvecan be considered to be the curve Rz-Cz to the left of 1000 cycles andcontinuing to the right of 1000 cycles into Ri-Cl. The curve labeledResponse inversely proportional to frequency is the theoreticalcompensation curve desired which will never reach the zero audiofrequency ordinate line. The actual cornpensation curve will be as shownat Rz-Cz. The curve Rz--Cz fails to conform with the theoretical curveat the very low frequencies, because of the interfering effect of theresistor Ra. 'I'he lowest frequency for which overall compensation isprovided is 400 cycles. Rs should have a very high value of resistancethereby to allow Rz--Cz to operate theoretically perfect. Hence, Rs isnot a part of the compensation network R2C2.

The modulation signal voltage output of tube 2' is preferably applied toan audio amplifier having a substantially fiat frequency responsecharacteristic. to insure presentation of the modulation signal energyto the reproducer with a flat audio response. Of course, any well-knowntype of flat gain amplier may be employed. It is to be clearlyunderstood that the present invention is not limited to the specificaudio amplifier system shown in Fig. 1 coupled to the output terminalsof the compensation network R2-C2. However, since I have found thespecific audio amplifier shown of desirable characteristics the presentclisclosure includes such specific circuit.

The audio voltage at the plate end of resistor R2 is applied to theinput grid of an audio amplier 4. The construction of the latter is ofgenerally well-known form. The amplifier stage including tube 4 is ofthe conventional resistancecondenser coupled type. Degenerative audiovoltage feedback is provided to the cathode of tube 4 by connecting thelower end of the bypassed cathode biasing resistor to the junction of apair of series-arranged resistors 6 and 1. These resistors are arrangedin the cathode circuit of power output tube I4. 'Ihe feedback lead 8connects the lower end of bias resistor 5 to ground through theresistor 1. The amplified audio voltage developed across the plateresistor 9 is applied to the input grid of the following audio ampliertube I0. The cathode of tube I0 is connected to ground through anunbypassed resistor I I.

Tubes 4 and I0 may be of the 6.15 type, although other types of tubessuitable for the particular purpose may be employed. The plate of tubeI0 is connected to a positive potential point through a series pathconsisting of resistor I2 and high frequency equalizing coil I3. Theaudio voltage developed across the plate load of tube I0 is applied tothe input grid of the power output tube I4 by means of a direct currentblocking condenser I5. The input grid of tube I 4 is connected to apositive potential point by means of a resistor I6. Assuming that thescreen and plate of the output tube I4 are connected to a +250y voltpoint, the resistor I6 may be connected to a voltage point ofapproximately +85 volts. The tube I4 may be of the GAG? type, althoughhere again The purpose of the latter characteristic is any otherwell-knownform of tube may be used.

The tube I4 is a so-called cathode follower" amplifier tube. That is tosay resistors Iiy .and 1 are unbypassed, and the load circuit is coupleddirectly across the cathode resistive load 6, l. 'Ihis is accomplishedby connecting the primary circuit of audio output transformer 20 inshunt across resistors 6 and l. A direct current blocking condenser 2lis connected in series with the primary winding of transformer 29. Thesecondary winding of the audio output transformer may be connected toany desired load, such as a loud speaker. The advantages of using thecathode follower amplifier I4 are the low distortion obtained in theoutput, and the constant low value of drive impedance at the transformeroutput terminals.

Actual experimental use with an audio amplier of the type describedprovided a reduction in distortion which was relatively great. Thefrequency response characteristic of the amplifier was substantiallyflat over a large range of useful audio and superaudio frequencies. Byway of specic illustration it is pointed out that the followingconstants may be employed in connection with the flat response audioamplifier of Fig. 1 it being understood that in the list of constantsthe symbol R designates resistance of the numbered resistor, the symbolC designates capacity of the numbered condenser and the symbol Ldesignates inductance of the numbered coil:

R 5=400 ohms R 6:4,000 ohms R 7:50 ohms R12=20,000 Ohms R 9=20,000 ohmsR11=500 ohms 021:1 microfarad L13=6.3 millihenry In Fig. 4 I have showna modification of the cascaded compensation networks wherein theyv arearranged in push-pull audio amplifier circuits. Thus, the compensationnetwork Ri-Ci is duplicated. The duplicate network is Ri-Ci. The pair ofcompensation networks R1-C1 and Ri-C'i is arranged in the output circuitof a balanced FM detector of the type disclosed by me in my pendingapplication Serial No. 418,927 filed November 13, 1941, U. S. PatentNumber 2,351,240 dated June 13, 1944. In this form of balanced FMdetector circuit the output load resistors 30 and 3| have the junctionthereof at ground potential. Accordingly, the ungrounded ends of theloadresistors 39 and 3l are respectively positive and negative inpolarity with respect to ground. When the applied signal energy has aninstantaneous mean frequency which is equal to the predeterminedreference frequency of the discriminator circuit, then the voltages atthe free ends of resistors 39 and 3! are of equal magnitude. Forfrequencies off the mean frequency the ends of resistors 30 and 3lassume potentials of different magnitude. The junction of thecompensation networks Ri--Ci and Ri-Ci is connected to ground.

The audio voltage developed in each of the compensation networks Ri-Ciand Ri-Ci is applied to a respective input grid of the pushpull audiofrequency amplifier tubes 40 and 4I through respective audio frequencycoupling condensers 49 and 4I. The tubes 40 and 4l are arranged inwell-known push-pull connections. The compensation network Rz-Cz isduplicated in the output circuit of tubes 40 and 4I, and the junction ofthe twonetworks Rz--Crl and Rz-Cz is establishedv at ground potential.In the case of this tube the resistor R2 of each compensation networkRz-Cz is fed with positive potential through a respective one ofresistors R3 and R's. It will be recognized that the arrangements aresubstantially the same as in the case of Fig. l, but that all circuitelements are duplicated for push-pull operation. The voltages developedacross each of the compensation resistors R2 are fed to a respective oneof the push-pull audio arnplier tubes 50 and 5| through respective audiocoupling condensers 5t' and 5I'. The control grid of tube 5D is returnedto Iground through the direct current return resistor Rs, while in thecase of the control grid of tube 5l there is used the return resistorR's. It will be understood that the plates of tubes 50 and 5I will beconnected in push-pull relation. Audio frequency voltage developed inthe plate circuits of tubes 5D and 5| will be fed to a push-pull audioamplifier having a nat frequency response characteristic.

By way of illustration, the following purely illustrative values aregiven for the circuit of Fig. si, it being understood, of course, thathere again the symbols R and C signify the resistance and capacityrespectively of the numbered resistors and condensers:

C1 and C1=0.004 microfarad C2 and C2=0.02 microfarad Ri and R1=100,000ohms R2 and R'2=20,000 ohms R3 and R3=100,000 ohms R4 and R'4=1 megohmRs and R'5=1 megohm Rao and Ra1=l00,000 ohms an output load resistoracross which is developed modulation signal voltage whose amplitude isan increasing function over a relatively wide range of modulationfrequencies, a modulation signal voltage amplifier, a first resistor andcondenser network in shunt with said load resistor and coupled to saidamplifier input, the ratio of resistive impedance to reactive impedanceof said network being suiliciently high to provide a4 frequency responseat said amplifier input which is inversely proportional to frequencyover a portion of said wide range above a predetermined median frequencythereof but which is relatively invariable below said median frequency,a second resistor and condenser network in shunt with said amplifieroutput, the constants of said second network being chosen to provide afrequency response characteristic which is inversely proportional tofrequency over that portion of said wide range below said medianfrequency but which is substantially invariable over said portion of therange above said median frequency whereby said F increasing amplitude iscompensated for over said wide range.

2. In an angle modulated carrier wave receiver of the type provided witha dernodulator having an 4output load resistor across which is developedmodulation signal voltage whose amplitude is an increasing function overa relatively wide rane of modulation frequencies, a modulation signalvoltage ampler, a first resistor and condenser network in shunt withsaid load resistor and coupled to said amplier input, the ratio ofresistive impedance to reactive impedance of said network beingsufficiently high to provide a. frequency response at said amplifierinput which is inversely proportional to frequency over a portion ofsaid wide range above a predetermined median frequency thereof but whichis relatively invariable below said median frequency, a second resistorand condenser network in shuntwith said amplifier output, the constantsof said second network being chosen to provide a frequency responsecharacteristic which is inversely proportional to frequency over thatportion of said wide range below said median frequency but which issubstantially invariable over said portion of the range above saidmedian frequency whereby said increasing amplitude is compensated forover said wide range and a non-frequency discriminatory modulationsignal voltage amplifier coupled to said second network,

3. In an angle modulated carrier wave receiver of the type provided witha demodulator having an output load resistor across which is developedmodulation signal voltage whose amplitude is an increasing function overa relatively wide range of modulation frequencies, a modulation signalvoltage amplifier, a rst resistor and condenser network in shunt withsaid load resistor and coupled to said amplifier input, the ratio ofresistive impedance to reactive impedance of said network beingsufficiently high to provide afrequency response at said amplier inputwhich is inversely proportional to frequency over a portion of said wide`range above a predetermined" median frequency thereof but which isrelatively invariable below said median frequency, a second resistor andcondenser network in shunt with said amplifier output, the constants ofsaid sec-. ond network being chosen to providea frequency responsecharacteristic-which is inversely proportional to frequency over thatportion of said wide range below said median frequency 4but whichissubstantially invariable-over said portion of the range above saidmedian frequency whereby said increasing amplitude is compensated forover said wide range, said first network having its resistor connectedin series vwith its condenser, said second network having its resistorconnected in series with its condenser, said wide range consisting ofthe audio frequency range, and said median frequency being substantially1000 cycles.

4. In combination with a source of audio frequency volt-age whoseamplitude increases proportionally with frequency over the audiofrequency range, an amplifier for the voltage, means operativelyassociated vwith the amplifier for compensating for said amplitudeincrease, said means comprising a rst voltage attenuation network,coupling said source to the amplifier input, whose constants are chosento provide increasing at tenuation of audio voltage for audiofrequencies of said range above a predetermined intermediate audiofrequency but relatively small attenuation below said intermediate audiofrequency value, a second voltage attenuation network, coupled to theamplifier output, whose constants are chosen toprovide relativelyuniform attenuation of said audio voltage for frequencies above'saidinterproportional to frequency over a portion of the audio range above arelatively low audio frequency but which response isrelativelyinvariable below that low frequency, and the secondcompensation network consisting of series resistance and capacity whoseconstants are chosen for rendering the low audio frequency response ofthe system inversely proportional to frequency for audio frequenciesbelow said low audio frequency and is relatively uniform above thelatter frequency.

BERTRAM TREVOR.

